1. Field of the Invention
This invention relates to a reception apparatus, a reception method and a program, and more particularly to a reception apparatus, a reception method and a program wherein a coefficient of an adaptive equalization filter used for demodulation of an OFDM signal can be readily produced with the OFDM time domain signal.
2. Description of the Related Art
A modulation method called orthogonal frequency division multiplexing (OFDM) method is known as a modulation method for ground wave digital broadcasting.
According to the OFDM method, a large number of orthogonal subcarriers are provided in a transmission band and data are allocated to the amplitude and the phase of the individual subcarriers and digitally modulated by phase shift keying (PSK) or Quadrature amplitude modulation (QAM).
The OFDM method is characterized in that, since the entire transmission band is divided by a large number of subcarriers, the bandwidth of one subcarrier is narrow and the transmission speed is low, but the total transmission speed is similar to that of conventional modulation methods. Further, the OFDM method is characterized in that the multi-path withstanding property can be improved by providing a guard interval hereinafter described.
Further, the OFDM system is characterized in that, since data are allocated to a plurality of subcarriers, a transmission circuit can be configured using an inverse fast Fourier transform (IFFT) mathematic operation circuit which carries out inverse Fourier transform upon modulation, and a reception circuit can be configured using a fast Fourier transform (FFT) mathematic operation circuit which carries out Fourier transform upon demodulation.
From such characteristics as described above, the OFDM method is frequently applied to ground wave digital broadcasting which is influenced strongly by a multi-path disturbance. As standards for ground wave digital broadcasting which adopt the OFDM method, such standards as, for example, the DVB-T (Digital Video Broadcasting-terrestrial), ISDB-T (Integrated Services Digital Broadcasting-Terrestrial) and ISDB-TSB are available.
FIG. 1 illustrates an OFDM symbol.
In the OFDM system, transmission of a signal is carried out in a unit called OFDM symbol.
Referring to FIG. 1, one OFDM symbol is composed of an effective symbol which is a signal interval within which IFFT is carried out upon transmission, and a guard interval (hereinafter referred to sometimes as GI) in which a waveform of part of a rear half of the effective symbol is copied. The GI is inserted to a position preceding to the effective symbol on the time axis.
According to the OFDM system, the GI is inserted so that interference of an OFDM symbol which occurs under a multi-path environment can be prevented.
A plurality of such OFDM symbols are gathered together to form one OFDM transmission frame. For example, according to the ISDB-T standards, one OFDM transmission frame is formed from 204 OFDM symbols. The insertion position of a pilot signal is defined with reference to a unit of the OFDM transmission frame.
In the OFDM system wherein a QAM type modulation system is used as a modulation system for subcarriers, since the subcarriers are influenced by multi-path interference or the like upon transmission, each subcarrier upon reception becomes different in amplitude and phase from the subcarrier upon transmission. Therefore, it is necessary to carry out signal equalization on the reception side so that the amplitude and the phase of a reception signal become equal to those of the transmission signal.
In the OFDM system, the transmission side inserts a pilot signal of a predetermined amplitude and a predetermined phase discretely into a transmission symbol. On the other hand, the reception side determines a frequency characteristic of the transmission line based on the amplitude and the phase of the pilot signal and equalizes the reception signal based on the determined characteristic of the transmission line.
The pilot signal used for calculation of a transmission line in this manner is called scattered pilot signal (hereinafter referred to as SP signal). FIG. 2 shows an arrangement pattern in OFDM symbols of an SP signal adopted by the DVB-T standards or the ISDE-T standards. In FIG. 2, the vertical direction is a time direction and the horizontal direction is a frequency direction.
FIG. 3 shows an example of a configuration of a conventional OFDM reception apparatus.
Referring to FIG. 3, the OFDM reception apparatus 1 shown includes a reception antenna 11, a tuner 12, a band-pass filter (BPF) 13, an analog to digital (A/D) conversion circuit 14, an orthogonal demodulation circuit 15, an FFT (Fast Fourier Transformation) circuit 16, an SP utilization equalization circuit 17 and an error correction circuit 18.
The reception antenna 11 receives a broadcasting wave broadcast from a broadcasting station and outputs an RF (radio frequency) signal to the tuner 12.
The tuner 12 includes a multiplication circuit 21 and a local oscillator 22, and frequency converts an RF signal received by the reception antenna 11 into an IF (intermediate frequency) signal and outputs the IF signal to the BPF 13.
The BPF 13 applies filtering to the IF signal supplied thereto from the tuner 12 and outputs a signal obtained by the filtering to the A/D conversion circuit 14.
The A/D conversion circuit 14 carries out A/D conversion for the signal supplied thereto from the BPF 13 and outputs a digital IF signal to the orthogonal demodulation circuit 15.
The orthogonal demodulation circuit 15 carries out orthogonal demodulation using a carrier signal of a predetermined frequency, that is, of a carrier frequency, to acquire an OFDM signal of a baseband from the IF signal supplied thereto from the A/D conversion circuit 14. This baseband OFDM signal is a signal in the time domain before FFT mathematic operation is applied.
A baseband OFDM signal before FFT mathematic operation is applied is hereinafter referred to as OFDM time domain signal. When the OFDM time domain signal is orthogonally demodulated, it becomes a complex signal which includes a real component (I-channel signal) and an imaginary component (Q-channel signal). The orthogonal demodulation circuit 15 outputs the time domain OFDM signal to the FFT circuit 16.
The FFT circuit 16 removes a signal within a range of a GI from the signal of one OFDM symbol to extract a signal within a range of an effective symbol length. The FFT circuit 16 carries out FFT mathematic operation for the extracted OFDM time domain signal to extract data orthogonally modulated in individual subcarriers.
The FFT circuit 16 outputs the OFDM signal representative of the extracted data to the SP utilization equalization circuit 17. The OFDM signal outputted from the FFT circuit 16 is a signal in the frequency domain after FFT mathematic operation is carried out. In the following description, an OFDM signal for which FFT mathematic operation has been carried out is referred to as OFDM frequency region signal.
The SP utilization equalization circuit 17 uses the SP signal arranged in such a manner as seen in FIG. 2 to calculate a transmission line characteristic of all of the subcarriers and compensates for a distortion of the OFDM frequency domain signal by the transmission line based on the calculated transmission line characteristics. The SP utilization equalization circuit 17 outputs the signal obtained by the compensation for the distortion by the transmission line as an equalization signal to the error correction circuit 18.
The error correction circuit 18 carries out a deinterleave process for the signal interleaved on the transmission side and further carries out such processes as depuncture, Viterbi decoding, spread signal removal and RS (Reed-Solomon) decoding. The error correction circuit 18 outputs data obtained by such various processes as decoded data to a circuit at the succeeding stage.
The OFDM system is characterized in that, by inserting a GI prior to an effective symbol, a demodulation process can be carried out without interference between symbols even in a multi-path environment within which a delay spread fits in a GI.
However, in such an environment wherein there is the possibility that long delay multi-paths may be produced as a single frequency network (SFN), the delay spread sometimes exceeds the GI. In this instance, inter-symbol interference or inter-carrier interference appears, and this deteriorates the reception performance significantly.
In order to solve this problem an OFDM reception apparatus 2 having such a configuration as shown in FIG. 4 and an OFDM reception apparatus 3 having such a configuration as shown in FIG. 5 have been proposed. In FIGS. 4 and 5, like elements to those of the OFDM reception apparatus 1 of FIG. 3 are denoted by like reference numerals.
The OFDM reception apparatus 2 of FIG. 4 includes, in addition to the components of the OFDM reception apparatus 1 shown in FIG. 3, an adaptive equalization filter 31 at a stage preceding to the FFT circuit 16. In the OFDM reception apparatus 2, the coefficient of the adaptive equalization filter 31 is adaptively controlled to remove multi-path components included in the OFDM time domain signal. The configuration of the adaptive equalization filter 31 is hereinafter described.
Meanwhile, the OFDM reception apparatus 3 includes, in addition to the components of the OFDM reception apparatus 1 shown in FIG. 3, an interference removing circuit 41 at a stage preceding to the FFT circuit 16.
The interference removing circuit 41 includes an adaptive equalization filter 51, a replica production apparatus 52 and a synthesis circuit 53. An OFDM time domain signal outputted from the orthogonal demodulation circuit 15 is inputted to the adaptive equalization filter 51 and the synthesis circuit 53.
The adaptive equalization filter 51 applies filtering to the OFDM time domain signal supplied from the orthogonal demodulation circuit 15 to remove multi-path components from the OFDM time domain signal and outputs the resulting OFDM time domain signal to the replica production apparatus 52.
The replica production apparatus 52 reproduces, based on the OFDM time domain signal supplied thereto from the adaptive equalization filter 51, the removed multi-path components and outputs a signal of the reproduced multi-path components to the synthesis circuit 53.
The synthesis circuit 53 removes inter-symbol interference components and inter-carrier interference components included in the FFT interval from the OFDM time domain signal supplied thereto from the orthogonal demodulation circuit 15 using the multi-path components reproduced by the replica production apparatus 52. The synthesis circuit 53 outputs the OFDM time domain signal from which the inter-symbol interference components and the inter-carrier interference components included in the FFT interval are removed to the FFT circuit 16.
A technique of reproducing a replica to remove interference components in this manner is disclosed in Japanese Patent Laid-Open No. 2007-6067.
Here, the adaptive equalization filter 31 shown in FIG. 4 is described.
FIG. 6 shows an example of a configuration of the adaptive equalization filter 31.
Referring to FIG. 6, the adaptive equalization filter 31 shown includes a variable coefficient filter 61, an SP extraction circuit 62, an IFFT circuit 63 and a main wave component removing circuit 64. The OFDM time domain signal outputted from the orthogonal demodulation circuit 15 is inputted to the variable coefficient filter 61, and the OFDM frequency domain signal outputted from the FFT circuit 16 is inputted to the SP extraction circuit 62.
The variable coefficient filter 61 applies filtering to the OFDM time domain signal supplied thereto from the orthogonal demodulation circuit 15 using a coefficient set based on a signal supplied from the main wave component removing circuit 64 to remove multi-path components included in the OFDM time domain signal. The variable coefficient filter 61 outputs the OFDM time domain signal from which the multi-path components are removed to the FFT circuit 16.
The SP extraction circuit 62 extracts an SP signal inserted at such a position as illustrated in FIG. 2 from the OFDM frequency domain signal supplied thereto from the FFT circuit 16 to remove modulation components to calculate a transmission path characteristic in the frequency domain. The SP extraction circuit 62 outputs the calculated transmission path characteristic to the IFFT circuit 63.
The IFFT circuit 63 carries out IFFT mathematic operation to convert the transmission path characteristic in the frequency domain into an impulse response characteristic of the transmission line in the time domain. The IFFT circuit 63 outputs the impulse response characteristic of the transmission line in the time domain to the main wave component removing circuit 64.
The main wave component removing circuit 64 removes a main wave component from the impulse response in the time domain calculated by the IFFT circuit 63 while leaving only the multi-path components and outputs a signal of the multi-path components to the variable coefficient filter 61. The variable coefficient filter 61 sets a coefficient corresponding to the amplitude and the phase of the multi-path components acquired by the main wave component removing circuit 64 to a tap corresponding to the delay time of the multi-path components to remove the multi-path components by filtering.
FIG. 7 shows an example of a configuration of the variable coefficient filter 61 shown in FIG. 6.
Referring to FIG. 7, the variable coefficient filter 61 includes a variable coefficient FIR (Finite-duration Impulse Response) filter 71 and a variable coefficient IIR (Infinite-duration Impulse Response) filter 72. Also a coefficient updating circuit not shown and some other circuits are provided in the variable coefficient filter 61. An OFDM time domain signal is inputted to the variable coefficient FIR filter 71.
The variable coefficient FIR filter 71 carries out filtering using a coefficient produced by the coefficient updating circuit not shown to remove or suppress multi-path (hereinafter referred to as pre-echo) components arriving earlier than the main wave.
The variable coefficient FIR filter 71 outputs a pre-echo equalized signal, which is an OFDM time domain signal whose pre-echo components are removed or suppressed, to the variable coefficient IIR filter 72. Since it is difficult to fully reduce pre-echo components, not only a signal from which pre-echo components are removed but also a signal within which pre-echo signals are suppressed are regarded as pre-echo equalized signal.
The variable coefficient IIR filter 72 includes a variable coefficient FIR filter 81 and a subtraction circuit 82 as shown in FIG. 7. The pre-echo equalized signal supplied from the variable coefficient FIR filter 71 is inputted to the subtraction circuit 82.
The variable coefficient FIR filter 81 applies filtering to the signal outputted from the subtraction circuit 82 using a coefficient produced by the coefficient updating circuit not shown and outputs a resulting signal to the subtraction circuit 82.
The subtraction circuit 82 subtracts the signal supplied thereto from the variable coefficient FIR filter 81 from the pre-echo equalized signal to remove multi-path (hereinafter referred to as post-echo) components arriving later than the main wave. Then, the subtraction circuit 82 outputs an equalization time domain signal obtained by removing the post-echo components. The equalization time domain signal outputted from the subtraction circuit 82 is inputted to the subtraction circuit 82 and also to the FFT circuit 16.
Also the adaptive equalization filter 51 shown in FIG. 5 has a configuration similar to that of the variable coefficient filter 61 described.
Like the OFDM reception apparatus 2 of FIG. 4 and the OFDM reception apparatus 3 of FIG. 5, an apparatus which removes multi-path components in the time domain includes a variable coefficient FIR filter and removes multi-path components by controlling the coefficient of the variable coefficient FIR filter.
Accordingly, if the coefficient is not appropriate, then multi-path components cannot be removed fully, and besides, a different multi-path component which has, as delay time, a time period equal to integral multiples of the delay time of an actually existing multi-path is added. This appears conspicuously particularly in such a case that updating of the coefficient cannot fully follow up the fluctuation by a Doppler effect or the like.
Since it is a practice to extract an SP signal from an OFDM frequency domain signal and estimate a transmission line characteristic based on the SP signal, such an algorithm as described above is an algorithm to which a delay profile is applied.
Meanwhile, as an adaptive algorithm which does not apply a delay profile, the least mean square (LMS) algorithm which minimizes a mean square error (MSE) is known. The LMS algorithm is a method which uses a known reference signal and is utilized most widely due to such characteristics that the adaptability performance is high and that the computation amount is small.
FIG. 8 shows an example of a circuit configuration to which the LMS algorithm is applied.
Referring to FIG. 8, the circuit shown includes a variable coefficient FIR filter 91 and a coefficient calculation circuit 92. Where the input signal is represented by x[k], the LMS algorithm is applied in order to reproduce a desired signal d[k] by filtering the input signal x[k] by means of the variable coefficient FIR filter 91.
The variable coefficient FIR filter 91 applies filtering to the input signal x[k] using a coefficient calculated by the coefficient calculation circuit 92 and outputs a resulting signal as an output signal y[k]. The output signal y[k] is outputted to the outside and also to the coefficient calculation circuit 92.
The coefficient calculation circuit 92 calculates a coefficient using the LMS algorithm and outputs the calculated coefficient to the variable coefficient FIR filter 91. Also the input signal x[k] is inputted to the coefficient calculation circuit 92.
In the example of FIG. 8, the coefficient calculation circuit 92 includes a subtraction circuit 101, a multiplication circuit 102, another multiplication circuit 103, an integration circuit 104, and a shift register 105.
The subtraction circuit 101 subtracts the desired signal d[k] from the output signal y[k] outputted from the variable coefficient FIR filter 91 to produce an error signal e[k]. The desired signal d[k] is a known signal, and the LMS algorithm can be applied where a known signal is available in this manner.
The multiplication circuit 102 is formed from a plurality of multipliers, which individually multiply the error signal e[k] and signals x[k−i] obtained by delaying the input signal x[k] by means of respective delay elements of the shift register 105. The multiplication circuit 102 outputs sample correlation values which are results of the multiplication by the individual multipliers to the multiplication circuit 103.
The multiplication circuit 103 is formed from a plurality of multipliers, which individually multiply the sample correlation values calculated by the multiplication circuit 102 by a step size μ and output results of the multiplication to the integration circuit 104.
The integration circuit 104 is formed from a plurality of integrators, which individually integrate the multiplication results of the multiplication circuit 103 to produce coefficients. The integration circuit 104 produces, for example, coefficients which cancel the sample correlation values representative of correlations of the error signals e[k] and the input signal x[k]. The integration circuit 104 sets the produced coefficients to respective taps of the variable coefficient FIR filter 91.
The processes described above which are carried out by the coefficient calculation circuit 92 can be represented by the following expressions. It is to be noted that γj[k] of the expression (1) given below represents a coefficient set to each tap of the variable coefficient FIR filter 91.
                              y          ⁡                      [            k            ]                          =                              ∑                          j              =              0                                      L              -              1                                ⁢                      γ            ⁢                                                  ⁢                          j              ⁡                              [                k                ]                                      *                          x              ⁡                              [                                  k                  -                  j                                ]                                                                        (        1        )                                          e          ⁡                      [            k            ]                          =                              y            ⁡                          [              k              ]                                -                      d            ⁡                          [              k              ]                                                          (        2        )                                          γ          ⁢                                          ⁢                      j            ⁡                          [                              k                +                1                            ]                                      =                              γ            ⁢                                                  ⁢                          j              ⁡                              [                k                ]                                              +                      μ            ·                          e              ⁡                              [                k                ]                                      ·                          x              ⁡                              [                                  k                  -                  j                                ]                                                                        (        3        )            As a variation to such an LMS algorithm as described above, also a Leaky-LMS algorithm is known wherein, when an integration process is carried out by the integration circuit 104, the integration process is carried out in a form wherein a Leak component is included in order to suppress the divergence of the coefficient or for a like object. The Leaky-LMS algorithm can be represented by the following expression (4) using a parameter λ:γj[k+1]=(1−λ)·γj[k]+μ·e[k]·x[k−j]  (4)Also the VSS (Variable Step Size)-LMS algorithm is known wherein the step size μ is variable in response to an error signal in order to improve the follow-up performance. As an updating algorithm for the step size μ, such an algorithm as represented by the following expressions (5) and (6) is known:μ[k+1]=(1−ε)·μ[k]+ν·∥e[k]∥^2  (5)μ[k+1]=(1−ε)·μ[k]+ν·ρ[k]^2ρ[k+1]=(1−η)·ρ[k]+η·e[k−1]·e[k]  (6)
As described above, in a circuit which uses the LMS algorithm, a sample correlation value between the error signal e[k] and the input signal x[k] is determined and updated in a direction in which it is canceled. Therefore, the coefficient finally converges to a value with which the error signal e[k] and the input signal x[k] have no correlation.